Showing posts with label controller. Show all posts
Showing posts with label controller. Show all posts
Saturday, December 21, 2013
Paraphase Tone Controller
As opposed to the widespread Baxandall circuit (dating back to 1952!) a ‘paraphrase’ tone control supplies a straight frequency response as long as the bass and treble controls are in the same position. This unique property makes the ‘paraphase’ configuration of interest if only treble or bass needs to be adjusted - it is not possible to adjust both at the same time! Essentially, it’s the difference in setting of the tone controls that determines the slope of the frequency response, and the degree of bass/treble correction. The circuit is simplicity itself, based on two networks C1-C2-C3/R9-R10-R11 and C5-C6-C7/R12-R13-R14.
Paraphase Tone Controller Circuit Picture:
The first is for the high frequencies (treble) response, the second, for the low frequencies (bass). The roll-off points have been selected, in combination with C4 and C8, for the sum of the two output signals to re-appear with a ‘straight’ frequency response again at the output. Roughly equal output levels from the networks are ensured by R6 = 7.15 k and R8 = 6.80 k. However, the operating principle requires the input signals to the two networks to be in anti-phase. For best operation the networks are driven by two buffers providing some extra gain.
Paraphase Tone Controller Circuit diagram:
The gain of IC1.D is slightly higher than that of IC1.C to ensure the overall response curve remains as flat as possible at equal settings of the tone controls. Because each network introduces a loss of about 1.72 (times), IC1.D and IC1.C first amplify the signal. The gain is set at about 8 (times) allowing input signal levels up to 1 V to pass the circuit at maximum gain and distortion-free. The gain also compensates the attenuation if you prefer to keep the tone controls at the mid positions for a straight response.
Parts and PCB layout:
Parts and PCB Layout
To audio fans, the circuit is rewarding to experiment with, especially in respect of the crossover point of the two networks. R3 and R4 determine the control range, which may be increased (within limits) by using lower resistor values here. The values shown ensure a tone control range of about 20 dB. IC1.B buffers the summed signal across R15. C9 removes any DC-offset voltage and R16 protects the output buffer from the effects of too high capacitive loads. R17, finally, keeps the output at 0 V. The choice of the quad opamp is relatively uncritical. Here the unassuming TL074 is used but you may even apply rail to rail opamps as long as they are stable at unity gain. Also, watch the supply voltage range. A simple circuit board was designed for the project. Linear-law potentiometers may be fitted directly onto the board. Two boards are required for a stereo application. The relevant connections on the boards are then wired to a stereo control potentiometer.
Specification:
- Current consumption (no signal) 8 mA
- Max. input signal 1 Veff (at max. gain)
- Gain at 20 Hz +13.1 dB max. –6.9 dB min.
- at 20 kHz +12.2 dB max. –7.6 dB min
- Gain (controls at mid position) 2.38 x
- Distortion (1 Veff, 1 kHz) 0.002% (B = 22kHz) 0.005% (B = 80 kHz)
COMPONENTS LIST
Resistors
R1-R4 = 10k
R5,R7 = 1k
R6 = 7k15
R8 = 6k80
R9,R10,R11 = 8k2
R12,R13,R14 = 2k2
R15 = 1M
R16 = 100R
R17 = 100k
P1,P2 = 100k preset or chassis-
mount control potentiometer, linear law
Capacitors
C1,C2,C3 = 47nF MKT, lead pitch 5mm
C4 = 68nF MKT, lead pitch 5mm
C5,C6,C7 = 10nF MKT, lead pitch 5mm
C8,C10,C11 = 100nF MKT, lead pitch 5mm
C9 = 2µF2 MKT, lead pitch 5mm or 7.5mm
Semiconductors
IC1 = TL074
Miscellaneous
K1,K2 = line socket, PCB mount, e.g.
T-709G (Monacor/Monarch)
R5,R7 = 1k
R6 = 7k15
R8 = 6k80
R9,R10,R11 = 8k2
R12,R13,R14 = 2k2
R15 = 1M
R16 = 100R
R17 = 100k
P1,P2 = 100k preset or chassis-
mount control potentiometer, linear law
Capacitors
C1,C2,C3 = 47nF MKT, lead pitch 5mm
C4 = 68nF MKT, lead pitch 5mm
C5,C6,C7 = 10nF MKT, lead pitch 5mm
C8,C10,C11 = 100nF MKT, lead pitch 5mm
C9 = 2µF2 MKT, lead pitch 5mm or 7.5mm
Semiconductors
IC1 = TL074
Miscellaneous
K1,K2 = line socket, PCB mount, e.g.
T-709G (Monacor/Monarch)
Source: http://www.ecircuitslab.com/2011/06/paraphase-tone-controller.html
Thursday, December 19, 2013
4 Amps Photovoltaic Solar Charge Controller
The use of solar photovoltaic (PV) energy sources is increasing due to global warming concerns on the one hand, and cost effectiveness on the other. Many engineers involved in power electronics find solar power tempting and then addictive due to the ‘green’ energy concept. The circuit discussed here handles up to 4 amps of current from a solar panel, which equates to about 75 watts of power. A charging algorithm called ‘pulse time modulation’ is introduced in this design. The current flow from the solar panel to the battery is controlled by an N-channel MOSFET, T1. This MOSFET does not require any heat sink to get rid of its heat, as its RD-S(on) rating is just 0.024 Ω.
Schottky diode D1 prevents the battery discharging into the solar panel at night, and also provides reverse polarity protection to the battery. In the schematic, the lines with a sort-of-red highlight indicate potentially higher current paths. The charge controller never draws current from the battery—it is fully powered by the solar panel. At night, the charge controller effectively goes to sleep. In daytime use, as soon as the solar panel produces enough current and voltage, it starts charging the battery. The battery terminal potential is divided by resistor R1 and trimpot P1.
The resulting voltage sets the charge state for the controller. The heart of the charge controller is IC1, a type TL431ACZ voltage reference device with an open-collector error amplifier. Here the battery sense voltage is constantly compared to the TL431’s internal reference voltage. As long as the level set on P1 is below the internal reference voltage, IC1 causes the MOSFET to conduct. As the battery begins to take up the charge, its terminal volt- age will increase. When the battery reaches the charge-state set point, the output of IC1 drops low to less than 2 volts and effectively turns off the MOSFET, stopping all current flow into the battery.
With T1 off, LED D2 also goes dark. There is no hysteresis path provided in the regulator IC. Consequently, as soon as the current to the battery stops, the output of IC1 remains low, preventing the MOSFET to conduct further even if the battery voltage drops. Lead-acid bat- tery chemistry demands float charging, so a very simple oscillator is implemented here to take care of this. Our oscillator exploits the negative resistance in transistors—first discovered by Leo Esaki and part of his studies into electron tunneling in solids, awarded with the Nobel Prize for Physics in 1973. In this implementation, a commonplace NPN transistor type 2SC1815 is used.
When the LED goes out, R4 charges a 22-μF capacitor (C1) until the voltage is high enough to cause the emitter-base junction of T2 to avalanche. At that point, the transistor turns on quickly and discharges the capacitor through R5. The voltage drop across R5 is sufficient to actuate T3, which in turn alters the reference voltage setting. Now the MOSFET again tries to charge the battery. As soon as the battery voltage reaches the charged level once more, the process repeats. A 2SC1815 transistor proved to work reliably in this circuit. Other transistors may be more temperamental—we suggest studying Esaki’s laureate work to find out why, but be cautioned that there are Heavy Mathematics Ahead.
As the battery becomes fully charged, the oscillator’s ‘on’ time shortens while the ‘off’ time remains long as determined by the timing components, R4 and C1. In effect, a pulse of current gets sent to the battery that will shorten over time. This charging algorithm may be dubbed Pulse Time Modulation. To adjust the circuit you’ll need a good digital voltmeter and a variable power supply. Adjust the supply to 14.9 V, that’s the 14.3 volts bat- tery setting plus approximately 0.6 volts across the Schottky diode.
Turn the trimpot until at a certain point the LED goes dark, this is the switch point, and the LED will start to flicker. You may have to try this adjustment more than once, as the closer you get the comparator to switch at exactly 14.3 V, the more accurate the charger will be. Disconnect the power supply from the charge controller and you are ready for the solar panel. The 14.3 V setting mentioned here should apply to most sealed and flooded-cell lead-acid batter- ies, but please check and verify the value with the manufacturer. Select the solar panel in such a way that its amps capability is within the safe charging limit of the battery you intend to use.
Resistors:
R1 = 15kΩ
R2,R3 = 3.3kΩ 1% R4 = 2.2MΩ
R5 = 1kΩ
P1 = 5kΩ preset
Capacitors:
C1 = 22μF 25V, radial
Semiconductors:
D1 = MBR1645G (ON Semiconductor) D2 = LED, 5mm
IC1 = TL431ACLP (Texas instruments)
T1 = IRFZ44NPBF (International Rectifier)
T2 = 2SC1815 (Toshiba) (device is marked: C1815)
T3 = BC547
Miscellaneous:
K1,K2 = 2-way PCB terminal block, lead pitch 5mm
Readmore...
Schottky diode D1 prevents the battery discharging into the solar panel at night, and also provides reverse polarity protection to the battery. In the schematic, the lines with a sort-of-red highlight indicate potentially higher current paths. The charge controller never draws current from the battery—it is fully powered by the solar panel. At night, the charge controller effectively goes to sleep. In daytime use, as soon as the solar panel produces enough current and voltage, it starts charging the battery. The battery terminal potential is divided by resistor R1 and trimpot P1.
 |
| 4 Amps Photovoltaic (Solar) Charge Controller Circuit Diagram |
With T1 off, LED D2 also goes dark. There is no hysteresis path provided in the regulator IC. Consequently, as soon as the current to the battery stops, the output of IC1 remains low, preventing the MOSFET to conduct further even if the battery voltage drops. Lead-acid bat- tery chemistry demands float charging, so a very simple oscillator is implemented here to take care of this. Our oscillator exploits the negative resistance in transistors—first discovered by Leo Esaki and part of his studies into electron tunneling in solids, awarded with the Nobel Prize for Physics in 1973. In this implementation, a commonplace NPN transistor type 2SC1815 is used.
When the LED goes out, R4 charges a 22-μF capacitor (C1) until the voltage is high enough to cause the emitter-base junction of T2 to avalanche. At that point, the transistor turns on quickly and discharges the capacitor through R5. The voltage drop across R5 is sufficient to actuate T3, which in turn alters the reference voltage setting. Now the MOSFET again tries to charge the battery. As soon as the battery voltage reaches the charged level once more, the process repeats. A 2SC1815 transistor proved to work reliably in this circuit. Other transistors may be more temperamental—we suggest studying Esaki’s laureate work to find out why, but be cautioned that there are Heavy Mathematics Ahead.
As the battery becomes fully charged, the oscillator’s ‘on’ time shortens while the ‘off’ time remains long as determined by the timing components, R4 and C1. In effect, a pulse of current gets sent to the battery that will shorten over time. This charging algorithm may be dubbed Pulse Time Modulation. To adjust the circuit you’ll need a good digital voltmeter and a variable power supply. Adjust the supply to 14.9 V, that’s the 14.3 volts bat- tery setting plus approximately 0.6 volts across the Schottky diode.
Turn the trimpot until at a certain point the LED goes dark, this is the switch point, and the LED will start to flicker. You may have to try this adjustment more than once, as the closer you get the comparator to switch at exactly 14.3 V, the more accurate the charger will be. Disconnect the power supply from the charge controller and you are ready for the solar panel. The 14.3 V setting mentioned here should apply to most sealed and flooded-cell lead-acid batter- ies, but please check and verify the value with the manufacturer. Select the solar panel in such a way that its amps capability is within the safe charging limit of the battery you intend to use.
Author: T. A. Babu (India - Elektor)
Resistors:
R1 = 15kΩ
R2,R3 = 3.3kΩ 1% R4 = 2.2MΩ
R5 = 1kΩ
P1 = 5kΩ preset
Capacitors:
C1 = 22μF 25V, radial
Semiconductors:
D1 = MBR1645G (ON Semiconductor) D2 = LED, 5mm
IC1 = TL431ACLP (Texas instruments)
T1 = IRFZ44NPBF (International Rectifier)
T2 = 2SC1815 (Toshiba) (device is marked: C1815)
T3 = BC547
Miscellaneous:
K1,K2 = 2-way PCB terminal block, lead pitch 5mm
Wednesday, September 4, 2013
Electronic Smart Heater Controller
Minuscule circuit of the electronic heater controller presented here is built around the renowned 3-Pin Integrated Temperature Sensor LM35 (IC1) from NSC. Besides, a popular Bi Mos Op-amp CA3140 (IC2) is used to sense the status of the temperature sensor IC1, which also controls a solid-state switch formed by a high power Triac BT136(T1). Resistive type electric heater at the output of T1 turns to ON and to OFF states as instructed by the control circuit.
This gadget can be used as an efficient and safe heater in living rooms, incubators, heavy electric/electronic instrument etc. Normally, when the temperature is below a set value (Decided by multi-turn preset pot P1), voltage at the inverting input (pin2) of IC1 is lower than the level at the non-inverting terminal (pin3). So, the comparator output (at pin 6) of IC1 goes high and T1 is triggered to supply mains power to the desired heater element.
Electronic Heater Controller Circuit diagram:
Note: CA3140 (IC2) is highly sensitive to electrostatic discharge (ESD). Please follow proper IC Handling Procedures.
When the temperature increases above the set value, say 50-60 degree centigrade, the inverting pin of IC1 also goes above the non-inverting pin and hence the comparator output falls. This stops triggering of T1 preventing the mains supply from reaching the heater element. Fortunately, the threshold value is user-controllable and can be set anywhere between 0 to 100 Degree centigrade.
The circuit works off stable 9Volt dc supply, which may be derived from the mains supply using a standard ac mains adaptor (100mA at 9V) or using a traditional capacitive voltage divider assembly. You can find such power circuits elsewhere in this website.
This gadget can be used as an efficient and safe heater in living rooms, incubators, heavy electric/electronic instrument etc. Normally, when the temperature is below a set value (Decided by multi-turn preset pot P1), voltage at the inverting input (pin2) of IC1 is lower than the level at the non-inverting terminal (pin3). So, the comparator output (at pin 6) of IC1 goes high and T1 is triggered to supply mains power to the desired heater element.
Electronic Heater Controller Circuit diagram:
Note: CA3140 (IC2) is highly sensitive to electrostatic discharge (ESD). Please follow proper IC Handling Procedures.
When the temperature increases above the set value, say 50-60 degree centigrade, the inverting pin of IC1 also goes above the non-inverting pin and hence the comparator output falls. This stops triggering of T1 preventing the mains supply from reaching the heater element. Fortunately, the threshold value is user-controllable and can be set anywhere between 0 to 100 Degree centigrade.
The circuit works off stable 9Volt dc supply, which may be derived from the mains supply using a standard ac mains adaptor (100mA at 9V) or using a traditional capacitive voltage divider assembly. You can find such power circuits elsewhere in this website.
Tuesday, September 3, 2013
MC14093 Pulse Width Modulation Controller
Using a MC14093 CMOS type IC, a quad 2-input NAND Schmitt trigger circuit can be designed a very simple pulse width modulation circuit controller electronic project . This pulse width modulation controller electronic project is very simple and require few external electronic parts . The speed is adjustable from 0-max. Max rpm is 2/3 the supply voltage. Supply voltage that can be used in this project can be between 3 and 18 volt .
Pulse Width Modulation Controller Schematic
MC14093 CMOS type IC can be directly interchanged with the MC14011 CMOS IC . Input pins 8, 9, 12, and 13, need to be connected to Gnd. or V+. Output pins 10 & 11 are left floating. Maximum current draw, with the components shown, is approximately 220 mA max using a small type motor. P1 potentiometer can be a multi-turn type if your needs are towards specific rpms. For Q1 transistor can be a IRF513, IRFZ42I IRF620 or IRF511 type .
To minimize RFI (Radio Frequency Interference), put a Ferrite Bead on the gate of Q1. A schottky diode of proper specs may be required for some motors which require faster switching .
Pulse Width Modulation Controller Schematic
To minimize RFI (Radio Frequency Interference), put a Ferrite Bead on the gate of Q1. A schottky diode of proper specs may be required for some motors which require faster switching .
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